Position sensorless control method in low-speed region of fault-tolerant permanent magnet motor system based on envelope detection and non-orthogonal phase-locked loop

ABSTRACT

In the position sensorless control method in low-speed region of the fault-tolerant permanent magnet motor system based on the envelope detection and the non-orthogonal phase-locked loop of the present disclosure, the position sensorless control of the motor is implemented by injecting the high-frequency voltage signals into any two non-faulty phase windings of the motor, extracting the high-frequency response currents of the high-frequency injected phases by the digital bandpass filter, calculating the differential mode inductances of the two phase windings through the envelope detecting and signal processing, and extracting the rotor position and rotational speed signals from the estimated two phase inductances through the non-orthogonal phase-locked loop. In addition, the controller of the present disclosure is small in size, high in accuracy, and high in reliability, which can effectively meet the performance requirements of the onboard electric actuators.

TECHNICAL FIELD

The present disclosure relates to the field of drive control technologyof a fault-tolerant permanent magnet motor, and in particular relates toa position sensorless control method in low-speed region of afault-tolerant permanent magnet motor system for a high-reliabilityonboard electric actuator based on envelope detection and anon-orthogonal phase-locked loop.

BACKGROUND

With increasing electrification of onboard equipment, themore-electric/all-electric aircraft gradually become a main developmentsubject of future aircraft due to significant technical advantages. Thetypical features of more-electric/all-electric aircraft are as follows:electric actuators have replaced hydraulic and pneumatic transmissionmechanisms in traditional aircraft; centralized hydraulic oil sourcesand pipelines across the entire aircraft body are reduced or completelyeliminated, realizing reduced aircraft weight, simplifying the secondaryenergy system structure of the aircraft, and greatly reducing theoperating cost of the aircraft. The more-electric/all-electric aircrafthas overcome the shortcomings of a traditional onboard hydraulic systemsuch as poor maintainability, low reliability, high noise, oil leakageand serious pollution, and improves the efficiency, maintainability,reliability and ground support capabilities of the aircraft. Electricactuators are widely applied in flight control systems, electricenvironmental control systems, electric fuel pump systems, enginecontrol systems, electric brake systems and landing gear.

The motor drive system is the core of an electric actuator, which has adecisive influence on performance of the entire actuation system. Tosatisfy the reliability requirement, the fault-tolerant permanent magnetmotor system is a primary choice for onboard electric actuator motorsystems due to advantages such as high power density, strong faulttolerance, good fault isolation, and high efficiency.

Position/speed detection is one of the key components for theclosed-loop control of fault-tolerant permanent magnet motor, which isassociated with the control performance of the control system. Theposition/speed detection of the motor is usually performed by a positionsensor. However, the use of the position sensor increases the hardwarecomplexity, volume and weight of the system, which also decreases thereliability of the system. Therefore, a position sensorless controlmethod is representative of an important research direction ofhigh-reliability motor drive systems.

The inventor once proposed a drive controller of a fault-tolerantpermanent magnet motor system for an onboard electric actuator and acontrol method for the fault-tolerant permanent magnet motor system in2019 (Chinese invention patent application No. ZL201910234708. 9),wherein a motor controller acquires current signals of respective phasewindings of the motor with Hall current sensors, and then motor speedand rotor position signals are estimated by a high-precision sensorlesscontrol method by using back-electromotive force (EMF) signals of twonon-faulty phase windings, thereby improving the accuracy andreliability of motor speed/position detection. In such control method,the back-EMFs of two phase windings are estimated by using a robustobserver model, and in turn, the control of the fault-tolerant permanentmagnet motor is achieved by extracting rotor position and rotationalspeed signals from the estimated back-EMF, therefore the control methodis appliable for fault-tolerant permanent magnet motors operating athigh speeds. However, although being also a sensorless control method,the control method can only be applicable to fault-tolerant permanentmagnet motors operating at high speeds, which can not be applicable tothe fault-tolerant permanent magnet motors operating at low speeds sincethe back-EMF method no longer works for the latter.

For the sensorless control of a permanent magnet synchronous motor(PMSM) in low-speed region, a high-frequency injection method is a mainstrategy. The conventional high-frequency injection method is based oncoordinate transformation and is dependent on the symmetry phasewindings of a motor. However, it is difficult to perform the coordinatetransformation when the fault-tolerant permanent magnet motor is runningin the fault condition, since the remaining non-faulty phase windingsare in an asymmetrical operation state. As a consequence, theconventional high-frequency injection method cannot realize the positionsensorless control of the fault-tolerant permanent magnet motor in faultoperation. In order to meet the reliability requirements of onboardelectric actuators, it is necessary to simplify the system structure andconduct research on sensorless control methods of fault-tolerantpermanent magnet motors in low-speed region. The present disclosureproposes a new position sensorless control method based on the envelopedetection and the non-orthogonal phase-locked loop, in which thehigh-frequency voltage signals are injected into any two non-faultyphase windings of the motor. The inductances of the two phase windingscan be estimated by the envelope detection, while the non-orthogonalphase-locked loop can be used for extracting the rotor position androtational speed information from the estimated two phase inductances.In the position sensorless control method according to the presentdisclosure, the high-frequency voltage signals are directly injectedinto any two non-faulty phase windings, and the method does not requireany coordinate transformation and does not rely on the symmetricaloperation of the phase windings, and thus is applicable to positionsensorless control of the fault-tolerant permanent magnet motors in bothnormal and faulty operation conditions.

SUMMARY

The present disclosure provides a highly-reliable sensorless controlmethod of fault-tolerant permanent magnet motor system in low-speedregion based on envelope detection and non-orthogonal phase-locked loop,which is appliable to the position sensorless control of the faulttolerant motor in fault operation condition. The essence of the presentdisclosure is to realize position sensorless control of thefault-tolerant permanent magnet motor in normal and faulty operations byinjecting the high-frequency voltage signals into any two non-faultyphase windings of the motor. The envelope detection is then proposed toestimate two phase inductance, while the non-orthogonal phase-lockedloop is proposed to extract the rotor position and rotational speedsignals in the estimated inductances.

In the position sensorless control method of the fault-tolerantpermanent magnet motor system of the present disclosure, thefault-tolerant permanent magnet motor system includes a fault-tolerantpermanent magnet motor, a signal detection circuit, an isolation drivecircuit, a fault-tolerant power drive circuit, a Digital SignalProcessor (DSP) system and a Field Programmable Gate Array (FPGA)system. The DSP system includes a speed loop controller, afault-tolerant controller and a sensorless resolving module, and isconfigured for the speed loop control, the calculation of thefault-tolerant current given instruction, the conditioning of theestimated inductance signal, and the resolving of the rotor position andspeed based on the differential mode inductance of two phase windings ofthe system. The FPGA system includes a current A/D sampling controlmodule, a current loop controller, a Pulse Width Modulation (PWM)generator module, a high-frequency signal generator, a digital bandpassfilter, a digital notch filter, an envelope detection algorithm module,a data transmission module and a fault diagnosis module, and isconfigured for the A/D sampling control, the current loop control, thehigh-frequency voltage signal injection and the corresponding signalprocessing for the sensorless control, the PWM signal generation and thefault diagnosis of the system.

The sensorless control method for the fault tolerant motor system inlowspeed region includes the following steps:

step 1: injecting a high-frequency sinusoidal voltage signal;

The high-frequency sinusoidal voltages are injected into any twonon-faulty phase windings of the motor by superimposing digitalhigh-frequency sinusoidal signals onto given voltages of twocorresponding phases acquired by a current loop, wherein thehigh-frequency voltage applied on the two phase windings is expressed asfollows:

$\begin{matrix}\left\{ {\begin{matrix}{U_{Ah} = {V_{h}\;{\sin\left( {{\omega_{h}t} + \varphi_{A}} \right)}}} \\{U_{Bh} = {V_{h}\;{\sin\left( {{\omega_{h}t} + \varphi_{B}} \right)}}}\end{matrix},} \right. & (1)\end{matrix}$

where U_(Ah), U_(Bh) respectively represent the high-frequency voltagesapplied on the terminals of the non-faulty phase windings A and B of themotor, V_(h) is the amplitude of the injected high-frequency voltage,ω_(h) is the frequency of the injected high-frequency voltage, andφ_(A), φ_(B) represent the initial phase of the injected high-frequencyvoltage;

according to the high-frequency approximate voltage model of thefault-tolerant permanent magnet motor, the high-frequency voltages areexpressed as follows:

$\begin{matrix}{\left\{ \begin{matrix}{U_{Ah} = {{L_{A}\left( \theta_{r} \right)}\frac{d\; i_{Ah}}{d\; t}}} \\{U_{Bh} = {{L_{B}\left( \theta_{r} \right)}\frac{d\; i_{Bh}}{d\; t}}}\end{matrix} \right.,} & (2)\end{matrix}$

where L_(A)(θ_(r)), L_(B)(θ_(r)) respectively represent theself-inductances of the winding A and the winding B, θ_(r) representsthe electrical angle of the rotor, and i_(Ah), i_(Bh) represent thehigh-frequency response currents of the winding A and the winding B,respectively;

from equations (1) and (2), the high-frequency response currents aresolved and expressed as follows:

$\begin{matrix}\left\{ {\begin{matrix}{i_{Ah} = {{F_{A}\left( {2\;\omega_{r}t} \right)}\sin\mspace{11mu}\left( {{\omega_{h}t} + {\varphi_{1}\left( \omega_{r} \right)}} \right)}} \\{i_{Bh} = {{F_{B}\left( {2\;\omega_{r}t} \right)}\sin\mspace{11mu}\left( {{\omega_{h}t} + {\varphi_{2}\left( \omega_{r} \right)}} \right)}}\end{matrix},} \right. & (3)\end{matrix}$

where F_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) respectively represent theamplitudes of the high-frequency response currents of phase A and phaseB, ω_(r) is the electrical angular velocity of the rotor, and theseparameters have the following approximate relationship with theself-inductances of the winding A and the winding B:

$\begin{matrix}{\left\{ \begin{matrix}{\frac{1}{F_{A}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{A}\left( {2\omega_{r}t} \right)}}} \\{\frac{1}{F_{B}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{B}\left( {2\omega_{r}t} \right)}}}\end{matrix} \right.;} & (4)\end{matrix}$

step 2: designing the digital notch filter and the digital bandpassfilter in the FPGA system according to the frequency of the injectedhigh-frequency voltage signal;

the designed digital notch filter is used to filter out thehigh-frequency current response components in the current loop, which isdesigned according to the frequency of the injected high-frequencyvoltage signal. Therefore, the digital notch filter has a stop band in avery small frequency band near ω_(h) and an amplitude attenuationgreater than 50 dB at ω_(h), which has almost no amplitude attenuationin other frequency bands and no phase lag;

the designed digital bandpass filter is used to extract thehigh-frequency response currents, which has a pass-band in a frequencyband near ω_(h) and a sufficiently high order for ensuring sufficientattenuation of the stop band;

both the digital notch filter and the digital bandpass filter arerequired to have the sufficient bandwidth, in which the samplingfrequency f_(s) should meet the following condition:

f _(s)>2f _(PWM)  (5),

where f_(PWM) is the PWM chopping frequency;

step 3: detecting the actual currents of the winding A and the winding Bby the signal detection circuit, and acquiring the high-frequencyresponse currents i_(Ah) and i_(Bh) of the winding A and the winding Bby the digital bandpass filter;

step 4: extracting the amplitude signals F_(A)(2ω_(r)t) andF_(B)(2ω_(r)t) of the high-frequency response current signals i_(Ah) andi_(Bh) by performing the envelope detection thereon respectively,specifically as follows:

in the digital system, the high-frequency response current signal inputin the envelope detection is expressed as follows:

f(nT _(s))=F(nT _(s))cos(ω_(h) nT _(s)+φ(nT _(s)))=(nT _(s))cos(ω_(h) nT_(s))−F _(Q)(nT _(s))sin(ω_(h) nT _(s))  (6),

where F(nT_(s)) represents the amplitude signal of the high-frequencyresponse current, F(nT_(s)) is the in-phase component, F_(Q)(nT_(s)) isthe quadrature component, n represents the nth sampling point, and T_(s)represents the sampling period;

a mixing process is first performed, in which the sine and cosinesignals with amplitude 2 and frequency ω_(h) generated by a digitallycontrolled oscillator are respectively multiplied with the inputhigh-frequency response current signal. Then a low-pass filteringprocess is performed, in which a product signal resulted in the mixingprocess is processed by the low-pass filter. After the mixing processand the low-pass filtering process, the following results are acquired:

$\begin{matrix}\left\{ {\begin{matrix}{I = {{{LP}{F\left( {{{f\left( {nT_{s}} \right)} \cdot 2}{\cos\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{I}\left( {nT_{s}} \right)}}} \\{Q = {{LP{F\ \left( {{{f\left( {nT_{s}} \right)} \cdot \ 2}{\sin\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{Q}\left( {nT_{s}} \right)}}}\end{matrix},} \right. & (7)\end{matrix}$

where I represents the in-phase component of the amplitude signalF(nT_(s)), Q represents the quadrature component of the amplitude signalF(nT_(s)), and LPF ( ) represents the process by a low-pass filter;

the processed signal is then subjected to square sum and square rootoperations and the following result is acquired:

√{square root over (I ² +Q ²)}=√{square root over (F(nT _(s))² +F_(Q)(nT _(s))²)}=F(nT _(s))  (8);

consequently, the amplitude information of the input signal is extractedafter a process with an envelope detection algorithm;

step 5: acquiring the estimated differential mode inductances

and

of the winding A and the winding B by conditioning the amplitude signalsF_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) of the high-frequency response currentsignals;

by the equation (4), the estimated self-inductances of the winding A andthe winding B are acquired by taking the inverse of F_(A)(2ω_(t)t) andF_(B)(2ω_(r)t) and multiplying the inverse by an appropriate gain,wherein the self-inductance includes a common mode component and adifferential mode component; then the estimated differential modeinductances are acquired by measuring the common mode component of theself-inductance and subtracting the common mode component from theself-inductance; and finally the estimated differential mode inductancesare properly filtered to filter out the interference signals;

step 6: resolving the estimated rotor electrical angular frequency

and the double estimated electrical angle 2

by inputting the estimated differential mode inductances

and

of the non-fault phase windings A and B into the non-orthogonalphase-locked loop, which is composed of a phase detector, a loop filterand a voltage-controlled oscillator;

wherein the estimated differential mode inductance signals

and

change at a double rotor angular frequency in one sine cycle, includingthe rotor speed and position information of the fault-tolerant permanentmagnet motor;

and

are non-orthogonal with a phase difference of 120°; the error relatedsignal between the estimated phase of the input signal and the actualphase of the input signal, i.e., the rotor position estimation errorrelated signal, is acquired by processing the two input estimatedinductance signals with the phase detector of the non-orthogonalphase-locked loop; and the following equation is satisfied for the phasedetector output PLL_(err) and the rotor position estimation error signalθ_(err):

$\begin{matrix}{{{PLL_{err}} = {{{\cdot {\cos\left( {{2\overset{\hat{}}{\theta}} - \frac{2\pi}{3}} \right)}} - {\cdot {\cos\left( {2\overset{\hat{}}{\theta}} \right)}}} = {{\frac{\sqrt{3}}{2}L{\sin\left( {2\theta_{err}} \right)}} \approx {\frac{\sqrt{3}}{2}{L \cdot 2}\theta_{err}}}}},} & (9)\end{matrix}$

wherein θ_(err) represents the error between the estimated rotorposition and the actual rotor position, L is a constant characterizingthe amplitude of the inductance signal; the loop filter of thenon-orthogonal phase-locked loop uses a PI controller, the rotorposition estimation error-related signal PLL_(err) input into the loopfilter is PI-modulated, and the double estimated rotor electricalangular frequency 2

and thus

are acquired after the modulating becomes stable; the double estimatedelectrical angle 2

is acquired by processing the 2

by an integrator of the voltage-controlled oscillator of thenon-orthogonal phase-locked loop; and the PI parameters of the PIcontroller of the loop filter are the design parameters of thenon-orthogonal phase-locked loop, and the PI parameters need thereasonable design and tuning for ensuring the stability and rapidity ofthe non-orthogonal phase-locked loop;

step 7: determining the polarity of the rotor permanent magnet of thefault-tolerant permanent magnet motor based on acceleration, resolvingthe estimated electrical angle

of the rotor, and performing the fixed compensation for the estimatedrotor position;

It is necessary to discriminate the polarity of the rotor permanentmagnet for determining

, due to the possible presence of a deviation of 180° between the valueacquired by dividing the double estimated electrical angle 2

resolved by the non-orthogonal phase-locked loop by 2 and the actualrotor position;

The discrimination of the polarity of the rotor permanent magnet isperformed only when the motor is started;

The fixed compensation is performed on the estimated rotor inductanceacquired from the phase-locked loop for ensuring control accuracy, sincethe phase of the estimated differential mode inductance signal lagsbehind its actual value due to the use of the filter.

Preferably, the fault-tolerant permanent magnet motor can be operated inthe failure mode of an open circuit fault and/or a short circuit faultfor a phase winding.

Preferably, determining the polarity of the rotor permanent magnet isperformed specifically as follows: before the motor is started, causingthe motor to operate on a trial basis according to the currentlyinitially rotor position estimated by the non-orthogonal phase-lockedloop for a short time (milliseconds); discriminating the polarity of thepermanent magnet by determining an acceleration direction of the rotorduring a short time, then determining that the estimated rotorelectrical angle is 2

/2 if the acceleration direction coincides with the excitation current,while determining that the estimated rotor electrical angle is

$\frac{2}{2} + \pi$

if the acceleration direction does not coincide with the excitationcurrent.

Preferably, the isolation drive circuit is composed of a gate isolationdrive chip and its peripheral circuits, which is used for the isolationbetween the strong electrical signal of the power driver and the weakelectrical signal of the digital controller for PWM control, and for thepower amplification of the PWM control signal generated by the FPGAsystem for driving and controlling turn-on and turn-off of powerMetal-Oxide-Semiconductor Field-Effect Transistors (MOSFETs).

Preferably, the fault-tolerant power drive circuit is implemented by anH-bridge power drive circuit based on power MOSFETs, and each phasewinding of the fault-tolerant permanent magnet motor is driven andpowered by a single H-bridge power drive circuit separately.

Preferably, the signal detection circuit includes a current sensor, asignal conditioning circuit and an A/D conversion circuit; the currentsensor is used to detect the phase current of each phase winding of thefault-tolerant permanent magnet motor and provides an output in the formof a voltage signal; the signal conditioning circuit is used to performa low-pass filtering process and a signal level conversion process onthe output from the current sensor, and output an analog signal to theA/D conversion circuit; and the A/D conversion circuit is used toconvert the analog signal into a digital signal and send the digitalsignal to the FPGA system. By injecting the high-frequency voltagesignals in the two non-faulty phase windings and processing thecorresponding phase current signals through the FPGA system and the DSPsystem, the differential mode inductance signals of the two non-faultyphase windings are finally resolved in the DSP system. Based on theobtained differential mode inductance signals, the high precisionsensorless resolving for the motor is completed, in which the rotorposition and speed of the motor in a low-speed region can be estimated,regardless of the fault state and the non-fault state. The given valueof the motor electromagnetic torque is acquired by the DSP systemthrough the calculation of the speed loop controller in the DSP systemaccording to the estimated motor speed and the control instructionsignal from the host computer. The given values of the currentinstruction of the non-faulty phase windings of the motor are calculatedby the fault-tolerant controller in the DSP system according to thegiven value of the motor electromagnetic torque instruction and thecurrent fault state of the motor system, and is then sent to the FPGAsystem; Finally, the FPGA system performs the current loop control andthe PWM generation.

Preferably, the DSP system is implemented by a floating-point high-speedDSP with a type number of TMS320F28335 with a main frequency of 150 MHzand a single-precision floating-point processing unit.

Preferably, the FPGA system includes a FPGA chip having a type number ofEP2C35F484 with a main frequency of 100 MHz, 33216 logics units, 105 M4Kmemory blocks, 35 multipliers, and 322 configurable I/O pins.

Preferably, the MOSFET devices in the fault-tolerant power drive circuithave a type number of IXTP90N075T2, which has the small on-state loss,small volume and weight, and high power density.

Preferably, the sensorless control method in low-speed region is appliedin the fault-tolerant permanent magnet motor system for electricactuators.

Beneficial effects of the present disclosure are as follows:

The present disclosure provides a high-precision position sensorlesscontrol method for the fault-tolerant permanent magnet motor systemoperating in the low-speed region based on high-frequency voltageinjection into two non-faulty phase windings. Through the real-timeobservation of differential mode inductance signals of any twonon-faulty phase windings, the rotor position and speed of the motor atstandstill or in low-speed operation can be estimated from the obtainedtwo phase inductances by the non-orthogonal phase-locked loop. Theposition sensorless control method does not rely on coordinatetransformation and does not require the symmetrical operation of phasewindings, and thus can be appliable for the high-precision control ofthe fault-tolerant permanent magnet motor operating at low-speeds inboth the fault state and the non-fault state, thereby improving theaccuracy and reliability of the motor position/speed detection.

The high-reliability position sensorless drive controller of thefault-tolerant permanent magnet motor system for electric actuatorsadopts the drive structure in which each phase winding is separatelypowered by an H-bridge inverter, and the core processor architecturebased on DSP and FPGA, thereby improving the fault isolation capabilityand fault-tolerant control performance of the system.

In addition, the power switch transistors of the drive controller have asmall conduction impedance, which effectively reduces the conductionloss; and the transistor is small in size, easy in installation, andhigh in power density.

The present disclosure provides a position sensorless control method ofthe fault-tolerant permanent magnet motor system operating in thelow-speed region for the electric actuator, which can improve the faultisolation capability and the control performance of the drive controllerof the fault-tolerant permanent magnet motor even in fault tolerantoperation condition by the proposed control strategy and structuralmodule design on the FPGA system. The control method of the presentdisclosure has strong reliability, high system efficiency, and smallsize, which can effectively meet the performance requirements of onboardelectric actuators.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of the overall structure of thefault-tolerant permanent magnet motor system for an onboard electricactuator according to the present disclosure;

FIG. 2 is a schematic diagram of component modules and functions of theFPGA system and the DSP system according to the present disclosure;

FIG. 3 is a schematic diagram of the structure of the fault-tolerantpower drive circuit according to the present disclosure;

FIG. 4 is a schematic diagram of the structure of the positionsensorless high-precision control algorithm based on envelope detectionand non-orthogonal phase-locked loop according to the presentdisclosure;

FIG. 5 is a schematic diagram of the specific algorithm of the envelopedetection algorithm according to the present disclosure; and

FIG. 6 is a schematic diagram of the specific algorithm of thenon-orthogonal phase-locked loop according to the present disclosure.

DETAILED DESCRIPTION

A specific embodiment of technical solutions of the present disclosurewill be described in detail below with reference to FIGS. 1-6.

FIG. 1 shows the block diagram of the high-reliability drive controllerof the fault-tolerant permanent magnet motor for the onboard electricactuator according to the present disclosure. The drive controllerincludes a floating-point high-speed DSP system, a large-scaleprogrammable logic gate array FPGA system, an isolation drive circuit, afault-tolerant power drive circuit and a signal detection circuit. TheDSP system is implemented by a floating-point high-speed DSP with a typenumber of TMS320F28335 with a main frequency of 150 MHz, and asingle-precision floating-point number processing unit. The FPGA systemis composed of an FPGA chip and its peripheral circuits, and the FPGAchip has a type number of EP2C35F484 (Cyclone II series, from ALTERA)with a main frequency of 100 MHz, 33216 logics units, 35 multipliers,322 configurable I/O pins. The MOSFET devices in the fault-tolerantpower drive circuit have a type number of IXTP90N075T2, which has smallconduction loss, small volume and weight, and high power density.

The high-reliability drive controller of the fault-tolerant permanentmagnet motor system for electric actuators is able to estimate the motorrotor position and speed information by: injecting the high-frequencyvoltage signals into two non-faulty phase windings of the motor,acquiring the corresponding current signals of the phase windings by thesignal detection circuit, resolving the current response signals of thehigh-frequency injection phases in the FPGA and the DSP through thesignal detection circuit, and accurately estimating the motor rotorposition and speed information through the high-precision sensorlesscontrol algorithm. The motor controller performs the followingprocesses: acquiring the given instruction of electromagnetic torquethrough the calculation of the speed loop controller according to theestimated speed and the speed instruction from a host computer;calculating the given current instruction of the non-faulty phase of themotor by the fault-tolerant controller according to the giveninstruction of electromagnetic torque, the estimated position and thefault state of the system; calculating the given voltages of thenon-fault phases by the current loop resolution according to the givencurrent instruction and the current feedback value detected by thesignal detection circuit, wherein the given voltages of the twonon-faulty phase windings as high-frequency injection phases arenecessarily superimposed with the high-frequency voltage signalgenerated by the high-frequency signal generator; and finally generatingthe PWM control instruction by the PWM generator module. The PWM controlinstruction is amplified in power by the isolation drive circuit and theamplified instruction is transmitted to the fault-tolerant power driverfor controlling the turn-on and turn-off of power switch transistorsthereof, so as to achieve the sensorless control of the fault-tolerantpermanent magnet motor and ensure the smooth operation thereof underfault or non-fault conditions.

As shown in FIG. 2, the main functions of the FPGA system include:controlling the current sampling by the A/D sampling control module;performing the fault diagnosis for the fault-tolerant permanent magnetmotor system; filtering the sampled current; acquiring the amplitudeinformation of the high-frequency currents by detecting the envelopes ofhigh-frequency response currents; acquiring the given voltageinstruction by the calculation of the current loop controller accordingto the given current instruction input from the DSP and the currentfeedback value measured by the A/D sampling control module; acquiringthe new given voltages of the corresponding phase windings bysuperimposing the high-frequency voltage signals generated by thehigh-frequency signal generator onto the given voltages of thecorresponding phase windings; and calculating the PWM control signals ofthe power switch transistors of respective phase windings by the PWMgenerator module according to the input given voltage signals ofrespective phases.

The DSP system is composed of a DSP chip and its peripheral circuits.The DSP chip is implemented by a 32-bit floating-point DSP with a typenumber of TMS320F28335 (from TI Inc. of US) with a main frequency of 150MHz, and a 32-bit floating-point processing unit. As shown in FIG. 2,the main functions of the DSP system include: estimating the real timerotor position and speed values of the motor in the low-speed region ofthe fault-tolerant permanent magnet motor system under fault andnon-fault conditions by resolution of the high-precision sensorlesscontroller according to the amplitude signals of the high-frequencyresponse currents resolved by the FPGA system; acquiring the givenvalues of electromagnetic torque by the calculation of the speed loopcontroller according to the speed control instruction from a hostcomputer and the estimated speed value; and acquiring the given currentinstruction of the non-faulty windings of the motor by the calculationof the fault-tolerant controller according to the given values of theelectromagnetic torque and the estimated rotor position, combined withthe failure mode given by the FPGA, and transmitting the given currentinstruction to the FPGA.

As shown in FIG. 3, the fault-tolerant power driver is composed of theH-bridge drive circuit, and each phase winding of the fault-tolerantpermanent magnet motor is driven and powered by one H-bridge drivecircuit separately. The power devices of the H-bridge drive circuit areimplemented by an N-channel enhancement MOSFET (from IXYS Inc. of US)with a type number of IXTP90N075T2 with the withstand voltage of 75V,the maximum current of 90 A, on-resistance of less than 10 mΩ, which issmall in size, low in loss and high in power density, and so on.

The overall sensorless control algorithm for the motor speed/positiondetection is shown in FIG. 4. The estimated differential mode inductancesignals of two non-faulty phase windings are acquired by: injecting thehigh-frequency voltages into the two non-faulty phase windings;detecting the currents of the two phase windings by the signal detectioncircuit; extracting the high-frequency response current components inthe detected currents by the bandpass filter; acquiring the amplitudesignals of the high-frequency response currents by performing theenvelope detection on the high-frequency response current components;and conditioning the amplitude signals. The differential mode inductancesignal contains the rotor speed/position information of the doublefrequency, and is input to the non-orthogonal phase-locked loop forresolution by which the estimated values of the double speed/positionare acquired. Finally, the estimated rotor speed/position of the motorcan be acquired after the discrimination of the polarity of the rotorpermanent magnet.

The isolation drive circuit is mainly composed of a gate isolation drivechip and peripheral circuits thereof. The isolation drive circuit hastwo functions: implementing the electrical isolation between the strongelectrical signal of the fault-tolerant power driver and the weakelectrical signal of the PWM controller, thereby improving the stabilityof the system; and performing the power amplification for the PWM signalgenerated by the FPGA system. The gate isolation drive chip isimplemented by an isolation-type high-precision half-bridge driver witha type number of ADuM7234 (from ADI Inc.), which isolates the highvoltage side and the low voltage side by the magnetic isolationtechnologies, and has the good isolation performance, a high operationfrequency up to 1 MHz, the strong anti-interference ability, the hightemperature resistance, and the small size.

The signal detection circuit includes a current sensor, a signalconditioning circuit and an A/D (analog-to-digital) converter. Thecurrent sensor is configured to detect the currents of respective phasesof the fault-tolerant permanent magnet motor. The current sensor isimplemented by a voltage-type Hall current sensor with a type number ofLTS 15-NP (from LEM Inc.), which has the fast response speed, the highprecision, the strong anti-interference ability, the good linearity, andthe small temperature drift. The signal conditioning circuit is composedof the operational amplifiers, the resistors and the capacitors, and isconfigured for applying the filtering and the level conversionprocessing on the signals acquired by the current sensor. The A/Dconverter is configured to convert the conditioned current analog signalinto a digital signal to be sent to the FPGA system. The A/D converterincludes an A/D converter chip implemented by an AD7606 chip (from ADIInc.), which is an 8-channel synchronous sampling data acquisition chipwith 14-bit conversion accuracy.

The position sensorless control method in low-speed region includes thefollowing steps:

step 1: injecting the high-frequency sinusoidal voltage signal;

the high-frequency sinusoidal voltages are injected into two non-faultyphase windings by superimposing the digital high-frequency sinusoidalsignals onto the given voltages of two corresponding phases acquired bythe current loop, wherein the high-frequency voltages applied on the twophase windings are expressed as follows:

$\begin{matrix}\left\{ {\begin{matrix}{U_{Ah} = {V_{h}{\sin\left( {{\omega_{h}t} + \varphi_{A}} \right)}}} \\{U_{Bh} = {V_{h}{\sin\left( {{\omega_{h}t} + \varphi_{B}} \right)}}}\end{matrix},} \right. & (1)\end{matrix}$

where U_(Ah), U_(Bh) respectively represent the high-frequency voltagesapplied on the terminals of non-faulty phase windings A and B of themotor, V_(h) is the amplitude of the injected high-frequency voltage,ω_(h) is the frequency of the injected high-frequency voltage, andφ_(A), φ_(B) represent the initial phase of the injected high-frequencyvoltage;

according to the high-frequency approximate voltage model of thefault-tolerant permanent magnet motor, the high-frequency voltages areexpressed as follows:

$\begin{matrix}{\left\{ \begin{matrix}{U_{Ah} = {{L_{A}\left( \theta_{r} \right)}\frac{d\; i_{Ah}}{d\; t}}} \\{U_{Bh} = {{L_{B}\left( \theta_{r} \right)}\frac{d\; i_{Bh}}{d\; t}}}\end{matrix} \right.,} & (2)\end{matrix}$

where L_(A)(θ_(r)), L_(B)(θ_(r)) respectively represent theself-inductances of the winding A and the winding B, θ_(r) representsthe electrical angle of the rotor, and i_(Ah), i_(Bh) represents thehigh-frequency response currents of the winding A and the winding B,respectively;

from equations (1) and (2), the high-frequency response currents aresolved and expressed as follows:

$\begin{matrix}\left\{ {\begin{matrix}{i_{Ah} = {{F_{A}\left( {2\;\omega_{r}t} \right)}\;\sin\;\left( {{\omega_{h}t} + {\varphi_{1}\left( \omega_{r} \right)}} \right)}} \\{i_{B\; h} = {{F_{B}\left( {2\;\omega_{r}t} \right)}\;\sin\;\left( {{\omega_{h}t} + {\varphi_{2}\left( \omega_{r} \right)}} \right)}}\end{matrix},} \right. & (3)\end{matrix}$

where F_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) respectively represent theamplitudes of the high-frequency response currents of phase A and phaseB, ω_(r) is the electrical angular velocity of the rotor, and theseparameters have the following approximate relationship with theself-inductances of the winding A and the winding B:

$\begin{matrix}{\left\{ \begin{matrix}{\frac{1}{F_{A}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{A}\left( {2\omega_{r}t} \right)}}} \\{\frac{1}{F_{B}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{B}\left( {2\omega_{r}t} \right)}}}\end{matrix} \right.;} & (4)\end{matrix}$

step 2: designing the digital notch filter and the digital bandpassfilter in the FPGA system according to the frequency of the injectedhigh-frequency voltage signal;

The designed digital notch filter is configured to filter out thehigh-frequency current response components in the current loop controlsignal and is designed according to the frequency of the injectedhigh-frequency voltage signal; and the digital notch filter has a stopband in a very small frequency band near ω_(h) and the amplitudeattenuation greater than 50 dB at ω_(h), and has almost no amplitudeattenuation in other frequency bands and no phase lag;

the designed digital bandpass filter is configured to extract thehigh-frequency response currents, and has a pass-band in the frequencyband near ω_(h) and a sufficiently high order for ensuring sufficientattenuation of the stop band;

both the digital notch filter and the digital bandpass filter have thesufficient bandwidth and have the sampling frequency f_(s) meeting thefollowing condition:

f _(s)>2f _(PWM)  (5),

where f_(PWM) is the PWM chopping frequency;

step 3: detecting the actual currents of the winding A and the winding Bby the signal detection circuit, and acquiring the high-frequencyresponse currents i_(Ah) and i_(Bh) of the winding A and the winding Bby the digital bandpass filter;

step 4: extracting the amplitude signals F_(A)(2ω_(r)t) andF_(B)(2ω_(r)t) of the high-frequency response current signals i_(Ah) andi_(Bh) by performing the envelope detection thereon respectively,specifically as follows:

in the digital system, the high-frequency response current signal inputin the envelope detection can be expressed as follows:

f(nT _(s))=F(nT _(s))cos(ω_(h) nT _(s)+φ(nT _(s)))=F _(I)(nT_(s))cos(ω_(h) nT _(s))−F _(Q)(nT _(s))sin(ω_(h) nT _(s))  (6),

where F(nT_(s)) represents the amplitude signal of the high-frequencyresponse current, F(nT_(s)) is the in-phase component, F_(Q)(nT_(s)) isthe quadrature component, n represents the nth sampling point, and T_(s)represents the sampling period;

a mixing process is performed, in which the sine and cosine signals withamplitude 2 and frequency ω_(h) generated by a digitally controlledoscillator are respectively multiplied with the input high-frequencyresponse current signal. A low-pass filtering process is then performed,in which a product signal resulted in the mixing process is processed bythe low-pass filter. After the mixing process and the low-pass filteringprocess, the following results are acquired:

$\begin{matrix}\left\{ {\begin{matrix}{I = {{{LP}{F\left( {{{f\left( {nT_{s}} \right)} \cdot 2}{\cos\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{I}\left( {nT_{s}} \right)}}} \\{Q = {{LP{F\ \left( {{{f\left( {nT_{s}} \right)} \cdot 2}{\sin\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{Q}\left( {nT_{s}} \right)}}}\end{matrix},} \right. & (7)\end{matrix}$

where I represents the in-phase component of the amplitude signalF(nT_(s)), Q represents the quadrature component of the amplitude signalF(nT_(s)), and LPF ( ) represents the process by a low-pass filter;

the processed signal is then subjected to the square sum and square rootoperations and the following result is acquired:

√{square root over (I ² +Q ²)}=√{square root over (F _(I)(nT _(s))² +F_(Q)(nT _(s))²)}=F(nT _(s))  (8);

consequently, the amplitude information of the input signal is extractedafter the process with the envelope detection algorithm;

step 5: acquiring the estimated differential mode inductances

and

of the winding A and the winding B by conditioning the amplitude signalsF_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) of the high-frequency response currentsignals;

by the equation (4), the estimated self-inductances of the winding A andthe winding B are acquired by taking the inverse of F_(A)(2ω_(r)t) andF_(B)(2ω_(r)t) and multiplying the inverse by an appropriate gain,wherein the self-inductance includes a common mode component and adifferential mode component; then the estimated differential modeinductances are acquired by measuring the common mode component of theself-inductance and subtracting the common mode component from theself-inductance; and finally the estimated differential mode inductancesare properly filtered to filter out the interference signals;

step 6: resolving the estimated rotor electrical angular frequency

and the double estimated electrical angle 2

by inputting the estimated differential mode inductances

and

of the non-fault phase windings A and B into the non-orthogonalphase-locked loop, which is composed of a phase detector, a loop filterand a voltage-controlled oscillator;

wherein the estimated differential mode inductance signals

and

change at a double rotor angular frequency in one sine cycle, includingthe speed and position information of the fault-tolerant permanentmagnet motor;

and

are non-orthogonal with a phase difference of 120°; the error relatedsignal between the estimated phase of the input signal and the actualphase of the input signal, i.e., the rotor position estimation errorrelated signal, can be acquired by processing the two input estimatedinductance signals with the phase detector of the non-orthogonalphase-locked loop; and the following equation is satisfied for the phasedetector output PLL_(err) and the rotor position estimation error signalθ_(err):

$\begin{matrix}{{{PLL_{err}} = {{{\cdot {\cos\left( {{2\overset{\hat{}}{\theta}} - \frac{2\pi}{3}} \right)}} - {\cdot {\cos\left( {2\overset{\hat{}}{\theta}} \right)}}} = {{\frac{\sqrt{3}}{2}L{\sin\left( {2\theta_{err}} \right)}} \approx {\frac{\sqrt{3}}{2}{L \cdot 2}\theta_{err}}}}},} & (9)\end{matrix}$

where θ_(err) represents the error between the estimated rotor positionand the actual rotor position, L is a constant characterizing theamplitude of the inductance signal; the loop filter of thenon-orthogonal phase-locked loop uses a PI controller, the rotorposition estimation error-related signal PLL_(err) input into the loopfilter is PI-modulated, and the double estimated rotor electricalangular frequency 2

and thus

are acquired after the modulating becomes stable; the double estimatedelectrical angle 2

is acquired by processing the 2

by an integrator of the voltage-controlled oscillator of thenon-orthogonal phase-locked loop; and the PI parameters of the PIcontroller of the loop filter are the design parameters of thenon-orthogonal phase-locked loop, and the PI parameters need thereasonable design and tuning for ensuring the stability and rapidity ofthe non-orthogonal phase-locked loop;

step 7: discriminating the polarity of the rotor permanent magnet of thefault-tolerant permanent magnet motor based on the acceleration,resolving the estimated electrical angle

of the rotor, and performing the fixed compensation for the estimatedrotor position;

it is necessary to discriminate the polarity of the rotor permanentmagnet for determining

, due to the possible presence of a deviation of 180° between the valueacquired by dividing the double estimated electrical angle 2

resolved by the non-orthogonal phase-locked loop by 2 and the actualrotor position;

the discrimination of the polarity of the rotor permanent magnet isperformed only when the motor is started; discriminating the polarity ofthe rotor permanent magnet is performed specifically as follows: beforethe motor is started, causing the motor to operate on a trial basisaccording to the currently initially rotor position estimated by thenon-orthogonal phase-locked loop for a short time (milliseconds);discriminating the polarity of the permanent magnet by determining theacceleration direction of the rotor during the short time, thendetermining that the estimated rotor electrical angle is 2

/2 if the acceleration direction coincides with an excitation current,while determining that the estimated rotor electrical angle is

2 + π

if the acceleration direction does not coincide with the excitationcurrent.

The fixed compensation is performed on the estimated rotor inductanceacquired from the phase-locked loop for ensuring the control accuracy,since the phase of the estimated differential mode inductance signallags behind the actual value due to the use of the filter.

The fault-tolerant permanent magnet motor can be operated in the faultmodes of an open circuit fault and/or a short circuit fault for a phasewinding.

In addition, it should be understood that although this specification isdescribed in accordance with embodiments, each embodiment includes morethan one independent technical solution. This description manner in thespecification is only for clarity, and those skilled in the art shouldconsider the specification as a whole. The technical solutions in theembodiments may also be appropriately combined to form other embodimentsthat may be understood by those skilled in the art.

1. A position sensorless control method in low-speed region of afault-tolerant permanent magnet motor system based on envelope detectionand a non-orthogonal phase-locked loop, wherein the fault-tolerantpermanent magnet motor system comprises a fault-tolerant permanentmagnet motor, a signal detection circuit, an isolation drive circuit, afault-tolerant power drive circuit, a DSP system and an FPGA system; theDSP system comprises a speed loop controller, a fault-tolerantcontroller and a sensorless resolving module, and the DSP system isconfigured for speed loop control, the calculation of a fault-tolerantcurrent given instruction, the conditioning of an estimated inductancesignal, and the resolving of a rotor position and a speed based ondifferential mode inductance of two phase windings of the system;wherein the FPGA system comprises a current A/D sampling control module,a current loop controller, a PWM generator module, a high-frequencysignal generator, a digital bandpass filter, a digital notch filter, anenvelope detection algorithm module, a data transmission module and afault diagnosis module, and the FPGA system is configured for the A/Dsampling control, the current loop control, the high-frequency voltagesignal injection and the corresponding signal processing for sensorlesscontrol, PWM signal generation and system fault diagnosis; thesensorless control method in low-speed region comprises the followingsteps: step 1: injecting the high-frequency sinusoidal voltage signals;the high-frequency sinusoidal voltages are injected into any twonon-faulty phase windings of the motor by superimposing digitalhigh-frequency sinusoidal signals onto given voltages of twocorresponding phases acquired by a current loop, wherein thehigh-frequency voltages applied on the two phase windings are expressedas follows: $\begin{matrix}\left\{ {\begin{matrix}{U_{Ah} = {V_{h}{\sin\left( {{\omega_{h}t} + \varphi_{A}} \right)}}} \\{U_{Bh} = {V_{h}{\sin\left( {{\omega_{h}t} + \varphi_{B}} \right)}}}\end{matrix},} \right. & (1)\end{matrix}$ where U_(Ah), U_(Bh) respectively represent thehigh-frequency voltages applied on terminals of non-faulty phasewindings A and B of the motor, V_(h) is the amplitude of the injectedhigh-frequency voltage, ω_(h) is the frequency of the injectedhigh-frequency voltage, and φ_(A), φ_(B) represent the initial phase ofthe injected high-frequency voltage; according to the high-frequencyapproximate voltage model of the fault-tolerant permanent magnet motor,the high-frequency voltages are expressed as follows: $\begin{matrix}{\left\{ \begin{matrix}{U_{Ah} = {{L_{A}\left( \theta_{r} \right)}\frac{d\; i_{Ah}}{d\; t}}} \\{U_{Bh} = {{L_{B}\left( \theta_{r} \right)}\frac{d\; i_{Bh}}{d\; t}}}\end{matrix} \right.,} & (2)\end{matrix}$ where L_(A)(θ_(r)), L_(B)(θ_(r)) respectively representthe self-inductances of the winding A and the winding B, θ_(r)represents the electrical angle of the rotor, and i_(Ah), i_(Bh)represent the high-frequency response currents of the winding A and thewinding B, respectively; from equations (1) and (2), the high-frequencyresponse currents are solved and expressed as follows: $\begin{matrix}\left\{ {\begin{matrix}{i_{Ah} = {{F_{A}\left( {2\;\omega_{r}t} \right)}\;\sin\;\left( {{\omega_{h}t} + {\varphi_{1}\left( \omega_{r} \right)}} \right)}} \\{i_{B\; h} = {{F_{B}\left( {2\;\omega_{r}t} \right)}\;\sin\;\left( {{\omega_{h}t} + {\varphi_{2}\left( \omega_{r} \right)}} \right)}}\end{matrix},} \right. & (3)\end{matrix}$ where F_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) respectivelyrepresent the amplitudes of the high-frequency response currents ofphase A and phase B, ω_(r) is the electrical angular velocity of therotor, and these parameters have the following approximate relationshipwith the self-inductances of the winding A and the winding B:$\begin{matrix}{\left\{ \begin{matrix}{\frac{1}{F_{A}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{A}\left( {2\omega_{r}t} \right)}}} \\{\frac{1}{F_{B}\left( {2\omega_{r}t} \right)} = {\frac{\omega_{h}}{V_{h}}{L_{B}\left( {2\omega_{r}t} \right)}}}\end{matrix} \right.;} & (4)\end{matrix}$ step 2: designing the digital notch filter and the digitalbandpass filter in the FPGA system according to the frequency of theinjected high-frequency voltage signal; the designed digital notchfilter is configured to filter out the high-frequency current responsecomponents in the current loop control signal and is designed accordingto the frequency of the injected high-frequency voltage signal; and thedigital notch filter has a stop band in a very small frequency band nearω_(h) and the amplitude attenuation greater than 50 dB at ω_(h), and hasalmost no amplitude attenuation in other frequency bands and no phaselag; the designed digital bandpass filter is configured to extract thehigh-frequency response currents, and has a pass-band in the frequencyband near ω_(h) and a sufficiently high order for ensuring sufficientattenuation of the stop band; both the digital notch filter and thedigital bandpass filter have the sufficient bandwidth and have thesampling frequency f_(s) meeting the following condition:f _(s)>2f _(PWM)  (5), where f_(PWM) is the PWM chopping frequency; step3: detecting the actual currents of the winding A and the winding B bythe signal detection circuit, and acquiring the high-frequency responsecurrents i_(Ah) and i_(Bh) of the winding A and the winding B by thedigital bandpass filter; step 4: extracting the amplitude signalsF_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) of the high-frequency response currentsignals i_(Ah) and i_(Bh) by performing the envelope detection thereonrespectively, specifically as follows: in a digital system, thehigh-frequency response current signal input in the envelope detectionis expressed as follows:f(nT _(s))=F(nT _(s))cos(ω_(h) nT _(s)+φ(nT _(s)))=F _(I)(nT_(s))cos(ω_(h) nT _(s))−F _(Q)(nT _(s))sin(ω_(h) nT _(s))  (6), whereF(nT_(s)) represents the amplitude signal of the high-frequency responsecurrent, F_(I)(nT_(s)) is the in-phase component, F_(Q)(nT_(s)) is thequadrature component, n represents the nth sampling point, and T_(s)represents the sampling period; a mixing process is performed, in whichthe sine and cosine signals with amplitude 2 and frequency ω_(h)generated by a digitally controlled oscillator are respectivelymultiplied with the input high-frequency response current signal, and alow-pass filtering process is performed, in which a product signalresulted in the mixing process is processed by a low-pass filter; afterthe mixing process and the low-pass filtering process, the followingresults are acquired: $\begin{matrix}\left\{ {\begin{matrix}{I = {{{LP}{F\left( {{{f\left( {nT_{s}} \right)} \cdot 2}{\cos\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{I}\left( {nT_{s}} \right)}}} \\{Q = {{LP{F\ \left( {{{f\left( {nT_{s}} \right)} \cdot 2}{\sin\left( {\omega_{h}nT_{s}} \right)}} \right)}} = {F_{Q}\left( {nT_{s}} \right)}}}\end{matrix},} \right. & (7)\end{matrix}$ where I represents the in-phase component of the amplitudesignal F(nT_(s)), Q represents the quadrature component of the amplitudesignal F(nT_(s)), and LPF ( ) represents the process by a low-passfilter; the processed signal is then subjected to square sum and squareroot operations and the following result is acquired:√{square root over (I ² +Q ²)}=√{square root over (F _(I)(nT _(s))² +F_(Q)(nT _(s))²)}=F(nT _(s))  (8); consequently, the amplitudeinformation of the input signal is extracted after the process with theenvelope detection algorithm; step 5: acquiring the estimateddifferential mode inductances

and

of the winding A and the winding B by conditioning the amplitude signalsF_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) of the high-frequency response currentsignals; by the equation (4), estimated the self-inductances of thewinding A and the winding B are acquired by taking the inverse ofF_(A)(2ω_(r)t) and F_(B)(2ω_(r)t) and multiplying the inverse by anappropriate gain, wherein the self-inductance comprises a common modecomponent and a differential mode component; then the estimateddifferential mode inductances are acquired by measuring the common modecomponent of the self-inductance and subtracting the common modecomponent from the self-inductance; and finally the estimateddifferential mode inductances are properly filtered to filter out theinterference signals; step 6: resolving the estimated rotor electricalangular frequency

and the double estimated electrical angle 2

by inputting the estimated differential mode inductances

and

of the non-fault phase windings A and B into the non-orthogonalphase-locked loop, which is composed of a phase detector, a loop filterand a voltage-controlled oscillator; wherein the estimated differentialmode inductance signals

and

change at the double rotor angular frequency in one sine cycle,including the speed and position information of the fault-tolerantpermanent magnet motor;

and

are non-orthogonal with a phase difference of 120°; the error relatedsignal between the estimated phase of the input signal and the actualphase of the input signal, i.e., the rotor position estimation errorrelated signal, is acquired by processing two input estimated inductancesignals with the phase detector of the non-orthogonal phase-locked loop;and the following equation is satisfied for the phase detector outputPLL_(err) and the rotor position estimation error signal θ_(err):$\begin{matrix}{{{{PL}L_{err}} = {{{{\cdot \cos}\left( {{2\overset{\hat{}}{\theta}} - \frac{2\pi}{3}} \right)} - {{\cdot \cos}\left( {2\overset{\hat{}}{\theta}} \right)}} = {{\frac{\sqrt{3}}{2}L{\sin\left( {2\theta_{err}} \right)}} \approx {\frac{\sqrt{3}}{2}{L \cdot 2}\theta_{err}}}}},} & (9)\end{matrix}$ where θ_(err) represents the error between the estimatedrotor position and the actual rotor position, L is a constantcharacterizing the amplitude of inductance signal; the loop filter ofthe non-orthogonal phase-locked loop uses a PI controller, the rotorposition estimation error-related signal PLL_(err) input into the loopfilter is PI-modulated, and the double estimated rotor electricalangular frequency 2

and thus

are acquired after the modulating becomes stable; the double estimatedelectrical angle 2

is acquired by processing the 2

by an integrator of the voltage-controlled oscillator of thenon-orthogonal phase-locked loop; and the PI parameters of the PIcontroller of the loop filter are the design parameters of thenon-orthogonal phase-locked loop, and the PI parameters need thereasonable design and tuning for ensuring the stability and rapidity ofthe non-orthogonal phase-locked loop; step 7: discriminating thepolarity of the rotor permanent magnet of the fault-tolerant permanentmagnet motor based on the acceleration, resolving the estimatedelectrical angle

of the rotor, and performing the fixed compensation for the estimatedrotor position; it is necessary to discriminate the polarity of therotor permanent magnet for determining

, due to the possible presence of a deviation of 180° between the valueacquired by dividing the double estimated electrical angle 2

resolved by the non-orthogonal phase-locked loop by 2 and the actualrotor position; the discrimination of the polarity of the rotorpermanent magnet is performed only when the motor is started; the fixedcompensation is performed on the estimated rotor inductance acquiredfrom the phase-locked loop for ensuring control accuracy, since thephase of the estimated differential mode inductance signal lags behindthe actual value due to the use of the filter.
 2. The control methodaccording to claim 1, wherein the fault-tolerant permanent magnet motorsystem can be operated in the fault mode of an open circuit fault and/ora short circuit fault for a phase winding.
 3. The control methodaccording to claim 1, wherein discriminating the polarity of the rotorpermanent magnet in step 7 is performed specifically as follows: beforethe motor is started, causing the motor to operate on a trial basisaccording to the currently initially rotor position estimated by thenon-orthogonal phase-locked loop for a short time, such as milliseconds;discriminating the polarity of the permanent magnet by determining theacceleration direction of the rotor during the short time, thendetermining that the estimated rotor electrical angle is 2

/2 if the acceleration direction coincides with an excitation current,while determining that the estimated rotor electrical angle is$\frac{2}{2} + \pi$ if the acceleration direction does not coincide withthe excitation current.
 4. The control method according to claim 1,wherein the isolation drive circuit is composed of a gate isolationdrive chip and peripheral circuits of the gate isolation drive chip,which is configured to implement isolation between the strong electricalsignal of the power driver and the weak electrical signal of the digitalcontroller for PWM control on the one hand, and to amplify the power ofthe PWM control signal generated by the FPGA system for driving andcontrolling the turn-on and turn-off of power MOSFETs on the other hand.5. The control method according to claim 1, wherein the fault-tolerantpower drive circuit is implemented by an H-bridge power drive circuitbased on the power MOSFETs, and each phase winding of the fault-tolerantpermanent magnet motor is driven and powered by a single H-bridge powerdrive circuit separately.
 6. The control method according to claim 1,wherein the signal detection circuit comprises a current sensor, asignal conditioning circuit and an A/D conversion circuit; the currentsensor detects the phase current of each phase winding of thefault-tolerant permanent magnet motor and provides an output in the formof a voltage signal; the signal conditioning circuit performs thelow-pass filtering process and the signal level conversion process onthe output from the current sensor, and output an analog signal to theA/D conversion circuit; and the A/D conversion circuit converts theanalog signal into a digital signal and send the digital signal to theFPGA system; and the differential mode inductance signals of the twonon-faulty phase windings are resolved in the DSP system finally byinjecting the high-frequency voltage signals in the two non-faulty phasewindings and processing the phase current signals of the correspondingwindings through the FPGA system and the DSP system, the high precisionsensorless resolving for the motor is completed thereby, and the rotorposition and speed in the low-speed region of the fault-tolerantpermanent magnet motor system in a fault state or in a non-fault stateare estimated in real time; the given value of the motor electromagnetictorque instruction is acquired by the DSP system through the calculationof the speed loop controller in the DSP system according to theestimated motor speed and the control instruction signal from the hostcomputer; the given value of the current instruction of the non-faultyphase winding of the motor is calculated by the fault-tolerantcontroller in the DSP system according to the given value of the motorelectromagnetic torque instruction and the current fault state of themotor system, and is sent to the FPGA system; and the FPGA systemperforms the current loop control and PWM generation.
 7. The controlmethod according to claim 1, wherein the DSP system is implemented by afloating-point high-speed DSP with a type number of TMS320F28335, a mainfrequency of 150 MHz, and a single-precision floating-point processingunit.
 8. The control method according to claim 1, wherein the FPGAsystem is composed of an FPGA chip and its peripheral circuits, and theFPGA chip has a type number of EP2C35F484 with a main frequency of 100MHz, 33216 logics units, 105 M4K memory blocks, 35 multipliers, and 322configurable I/O pins; the functions of the FPGA system include:controlling the current sampling by the A/D sampling control module;performing the fault diagnosis for the fault-tolerant permanent magnetmotor system; filtering the sampled current; acquiring the amplitudeinformation of the high-frequency currents by detecting the envelopes ofhigh-frequency response currents; acquiring the given voltageinstruction by the calculation of the current loop controller accordingto the given current instruction input from the DSP and the currentfeedback value measured by the A/D sampling control module; acquiringthe new given voltages of the corresponding phase windings bysuperimposing the high-frequency voltage signals generated by thehigh-frequency signal generator onto the given voltages of thecorresponding phase windings; and calculating the PWM control signals ofthe power switch transistors of respective phase windings by the PWMgenerator module according to the input given voltage signals ofrespective phases.
 9. The control method according to claim 1, whereinthe power MOSFET devices in the fault-tolerant power drive circuit havea type number of IXTP90N075T2, which has small conduction loss, smallvolume and weight, and high power density.
 10. The control methodaccording to claim 1, wherein the sensorless control method in low-speedregion can be applied for the fault-tolerant permanent magnet motorsystem of the electric actuators.
 11. The control method according toclaim 2 wherein discriminating the polarity of the rotor permanentmagnet in step 7 is performed specifically as follows: before the motoris started, causing the motor to operate on a trial basis according tothe currently initially rotor position estimated by the non-orthogonalphase-locked loop for a short time, such as milliseconds; discriminatingthe polarity of the permanent magnet by determining the accelerationdirection of the rotor during the short time, then determining that theestimated rotor electrical angle is 2

/2 if the acceleration direction coincides with an excitation current,while determining that the estimated rotor electrical angle is$\frac{2}{2} + \pi$ if the acceleration direction does not coincide withthe excitation current.
 12. The control method according to claim 2,wherein the isolation drive circuit is composed of a gate isolationdrive chip and peripheral circuits of the gate isolation drive chip,which is configured to implement isolation between the strong electricalsignal of the power driver and the weak electrical signal of the digitalcontroller for PWM control on the one hand, and to amplify the power ofthe PWM control signal generated by the FPGA system for driving andcontrolling the turn-on and turn-off of power MOSFETs on the other hand.13. The control method according to claim 3, wherein the isolation drivecircuit is composed of a gate isolation drive chip and peripheralcircuits of the gate isolation drive chip, which is configured toimplement isolation between the strong electrical signal of the powerdriver and the weak electrical signal of the digital controller for PWMcontrol on the one hand, and to amplify the power of the PWM controlsignal generated by the FPGA system for driving and controlling theturn-on and turn-off of power MOSFETs on the other hand.
 14. The controlmethod according to claim 2, wherein the fault-tolerant power drivecircuit is implemented by an H-bridge power drive circuit based on thepower MOSFETs, and each phase winding of the fault-tolerant permanentmagnet motor is driven and powered by a single H-bridge power drivecircuit separately.
 15. The control method according to claim 3, whereinthe fault-tolerant power drive circuit is implemented by an H-bridgepower drive circuit based on the power MOSFETs, and each phase windingof the fault-tolerant permanent magnet motor is driven and powered by asingle H-bridge power drive circuit separately.
 16. The control methodaccording to claim 4, wherein the fault-tolerant power drive circuit isimplemented by an H-bridge power drive circuit based on the powerMOSFETs, and each phase winding of the fault-tolerant permanent magnetmotor is driven and powered by a single H-bridge power drive circuitseparately.
 17. The control method according to claim 2, wherein thesignal detection circuit comprises a current sensor, a signalconditioning circuit and an A/D conversion circuit; the current sensordetects the phase current of each phase winding of the fault-tolerantpermanent magnet motor and provides an output in the form of a voltagesignal; the signal conditioning circuit performs the low-pass filteringprocess and the signal level conversion process on the output from thecurrent sensor, and output an analog signal to the A/D conversioncircuit; and the A/D conversion circuit converts the analog signal intoa digital signal and send the digital signal to the FPGA system; and thedifferential mode inductance signals of the two non-faulty phasewindings are resolved in the DSP system finally by injecting thehigh-frequency voltage signals in the two non-faulty phase windings andprocessing the phase current signals of the corresponding windingsthrough the FPGA system and the DSP system, the high precisionsensorless resolving for the motor is completed thereby, and the rotorposition and speed in the low-speed region of the fault-tolerantpermanent magnet motor system in a fault state or in a non-fault stateare estimated in real time; the given value of the motor electromagnetictorque instruction is acquired by the DSP system through the calculationof the speed loop controller in the DSP system according to theestimated motor speed and the control instruction signal from the hostcomputer; the given value of the current instruction of the non-faultyphase winding of the motor is calculated by the fault-tolerantcontroller in the DSP system according to the given value of the motorelectromagnetic torque instruction and the current fault state of themotor system, and is sent to the FPGA system; and the FPGA systemperforms the current loop control and PWM generation.
 18. The controlmethod according to claim 3, wherein the signal detection circuitcomprises a current sensor, a signal conditioning circuit and an A/Dconversion circuit; the current sensor detects the phase current of eachphase winding of the fault-tolerant permanent magnet motor and providesan output in the form of a voltage signal; the signal conditioningcircuit performs the low-pass filtering process and the signal levelconversion process on the output from the current sensor, and output ananalog signal to the A/D conversion circuit; and the A/D conversioncircuit converts the analog signal into a digital signal and send thedigital signal to the FPGA system; and the differential mode inductancesignals of the two non-faulty phase windings are resolved in the DSPsystem finally by injecting the high-frequency voltage signals in thetwo non-faulty phase windings and processing the phase current signalsof the corresponding windings through the FPGA system and the DSPsystem, the high precision sensorless resolving for the motor iscompleted thereby, and the rotor position and speed in the low-speedregion of the fault-tolerant permanent magnet motor system in a faultstate or in a non-fault state are estimated in real time; the givenvalue of the motor electromagnetic torque instruction is acquired by theDSP system through the calculation of the speed loop controller in theDSP system according to the estimated motor speed and the controlinstruction signal from the host computer; the given value of thecurrent instruction of the non-faulty phase winding of the motor iscalculated by the fault-tolerant controller in the DSP system accordingto the given value of the motor electromagnetic torque instruction andthe current fault state of the motor system, and is sent to the FPGAsystem; and the FPGA system performs the current loop control and PWMgeneration.
 19. The control method according to claim 4, wherein thesignal detection circuit comprises a current sensor, a signalconditioning circuit and an A/D conversion circuit; the current sensordetects the phase current of each phase winding of the fault-tolerantpermanent magnet motor and provides an output in the form of a voltagesignal; the signal conditioning circuit performs the low-pass filteringprocess and the signal level conversion process on the output from thecurrent sensor, and output an analog signal to the A/D conversioncircuit; and the A/D conversion circuit converts the analog signal intoa digital signal and send the digital signal to the FPGA system; and thedifferential mode inductance signals of the two non-faulty phasewindings are resolved in the DSP system finally by injecting thehigh-frequency voltage signals in the two non-faulty phase windings andprocessing the phase current signals of the corresponding windingsthrough the FPGA system and the DSP system, the high precisionsensorless resolving for the motor is completed thereby, and the rotorposition and speed in the low-speed region of the fault-tolerantpermanent magnet motor system in a fault state or in a non-fault stateare estimated in real time; the given value of the motor electromagnetictorque instruction is acquired by the DSP system through the calculationof the speed loop controller in the DSP system according to theestimated motor speed and the control instruction signal from the hostcomputer; the given value of the current instruction of the non-faultyphase winding of the motor is calculated by the fault-tolerantcontroller in the DSP system according to the given value of the motorelectromagnetic torque instruction and the current fault state of themotor system, and is sent to the FPGA system; and the FPGA systemperforms the current loop control and PWM generation.
 20. The controlmethod according to claim 5, wherein the signal detection circuitcomprises a current sensor, a signal conditioning circuit and an A/Dconversion circuit; the current sensor detects the phase current of eachphase winding of the fault-tolerant permanent magnet motor and providesan output in the form of a voltage signal; the signal conditioningcircuit performs the low-pass filtering process and the signal levelconversion process on the output from the current sensor, and output ananalog signal to the A/D conversion circuit; and the A/D conversioncircuit converts the analog signal into a digital signal and send thedigital signal to the FPGA system; and the differential mode inductancesignals of the two non-faulty phase windings are resolved in the DSPsystem finally by injecting the high-frequency voltage signals in thetwo non-faulty phase windings and processing the phase current signalsof the corresponding windings through the FPGA system and the DSPsystem, the high precision sensorless resolving for the motor iscompleted thereby, and the rotor position and speed in the low-speedregion of the fault-tolerant permanent magnet motor system in a faultstate or in a non-fault state are estimated in real time; the givenvalue of the motor electromagnetic torque instruction is acquired by theDSP system through the calculation of the speed loop controller in theDSP system according to the estimated motor speed and the controlinstruction signal from the host computer; the given value of thecurrent instruction of the non-faulty phase winding of the motor iscalculated by the fault-tolerant controller in the DSP system accordingto the given value of the motor electromagnetic torque instruction andthe current fault state of the motor system, and is sent to the FPGAsystem; and the FPGA system performs the current loop control and PWMgeneration.